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1970s Design Indulgence

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Graham Slee View Drop Down
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Post Options Post Options   Thanks (0) Thanks(0)   Quote Graham Slee Quote  Post ReplyReply Direct Link To This Post Posted: 13 Jan 2019 at 5:59am
Hi Bruce, answering your questions...

1. Yes. Within constraints that's correct.

For example, increasing the VAS local feedback capacitor (Cdom) any further than necessary will not improve phase any further, because being in a NFB loop with substantial transistor base-emitter capacitances (in the output stage), it has already been subjected to phase shift. This is like what happens with an op-amp driving a capacitive load. The only way around that is to drive the output stage with more VAS current, leading to an even-hotter VAS transistor, or making the output stage a triple instead of a double. Even so, we will be faced with the transition frequency of the last transistor.

Decreasing the value steepens the fall-off, increasing phase shift, which might still be considered stable, but anything less than 90 degrees (steeper than 20dB/decade) causes ringing. The output capacitor starts to shift from being "purely" capacitive to becoming inductive at high frequencies (10kHz upwards for example), and excessive ringing can heat it up, which eats into its lifetime, "wearing it out" little by little.

Here bypassing might help, but not to get more HF to pass for subjective improvement/difference, but simply to add some protection to the output capacitor. This might be facilitated by joining the junction of the output capacitor and the output inductor with the junction of the 4u7 capacitor and the 100 ohm resistor. I shall have to give it a try.

2. In theory the input filter should "slow" the signal such that it cannot exceed the available slew-rate, and this reduces the chance of ringing. Getting it to work in practice is the difficult/impossible bit: The value is calculated from the input series resistor combined with the output resistance of the source. 2k2 is the maximum "standards" value from a solid-state source, but what about a valve preamp with a 100k volume pot on its output? Having realised that since my post, I decided against putting much design emphasis on it, and simply seeing it as EMI blocking.

3. The inductor serves the purpose of isolating the output from complex loads (in a similar way to using a 100 ohm resistor on an op-amp output).

The Zobel network, operating before the output inductor, provides a fixed and simpler value of loading at high frequency. It is mainly required for the purpose of stopping the emitter follower pair (the upper part of the output stage, T4 and T6) going into "Colpitt's oscillation". If it operates before the 0dB crossing it adds to the phase shift making ringing worse. If it operates too far beyond the 0dB crossing it allows the onset of "Colpitt's oscillation". The effect can be seen as a slight bellying out (toward the right and upwards) of the gain curve below the 0dB crossing, which at first looks quite benign because the simulator shows improved phase margin. But this is a trap.

Tandberg (and I think Aiwa did it too) got away with it because of the poor HF extension of the germanium output devices which made the difficulties highlighted above less of an issue. Progress has led to improvements which have in-turn led to problems...

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Post Options Post Options   Thanks (0) Thanks(0)   Quote BAK Quote  Post ReplyReply Direct Link To This Post Posted: 17 Jan 2019 at 4:43pm
Originally posted by Graham Slee Graham Slee wrote:

Hi Bruce, answering your questions...

2. In theory the input filter should "slow" the signal such that it cannot exceed the available slew-rate, and this reduces the chance of ringing. Getting it to work in practice is the difficult/impossible bit: The value is calculated from the input series resistor combined with the output resistance of the source. 2k2 is the maximum "standards" value from a solid-state source, but what about a valve preamp with a 100k volume pot on its output? Having realised that since my post, I decided against putting much design emphasis on it, and simply seeing it as EMI blocking.

Could one consider designing the input filter to block RF intrusion would have an added benefit of reducing the chance of ringing due to over-running the slew-rate?
(Thinking... If the RF filter cap was large enough to block EMI, it should be big enough to slow the input, too.)
Another balancing act... electronic design is full of compromises.


Edited by BAK - 17 Jan 2019 at 4:45pm
Bruce
AT-14SA, Pickering XV-15/625, Technics SL-1600MK2, Reflex M, Lautus, Technics SH-8066, Dynaco ST120a, Eminence Beta 8A in custom cabs;; Using Majestic DAC
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Post Options Post Options   Thanks (0) Thanks(0)   Quote Graham Slee Quote  Post ReplyReply Direct Link To This Post Posted: Yesterday at 6:49am
Finishing The Job?

(power amp section)

So little is written about this power amplifier circuit configuration that it feels like I'm finishing the job started by HC Lin, Tobey and Dinsdale - filling-in the many blanks they left behind.

Take on a slightly different configuration and there's a wealth of published information, and if you decide to go the common route of the long tailed pair input stage you don't even need to be an electronics design engineer! You just copy one of the myriad of published designs.

DC Stability

There is however one article which covers the class-A voltage amplifier to some degree with fig. 11 page 41 here: https://www.americanradiohistory.com/Archive-Wireless-World/70s/Wireless-World-1971-01.pdf - being the closest thing to what I have.

The limited information here helped jog my memory about proportioning supply voltage for DC stability. It helped me understand a "rule of thumb" gleaned from my research nearly 43 years ago.

It is an ambiguous rule of thumb that should have read that the emitter voltage of T2 should be 1/10th the supply. But it discussed a 9V supply and rounded up the emitter voltage to 1V, so the memory might interpret that in any such design only 1V is required. In this design I have proven for myself that 1/10th or thereabouts is the requirement, and the 3-4 volts I had made settling the DC a "regular task", which means not stable.

This design is so close to that of my Proprius that I expected it to sound almost identical, but it took on a "roughness", and I can only put that down to a change in distortion between the test bench and the listening room, and the only difference is in the mains supply voltage: test bench is adjusted to 230V (by means of a giant 'variac'); and listening room is on 245 - 254 volts depending on substation loading.

Rather than recalculating, it appeared logical to simply use the same values as in the Proprius for collector load and emitter, which contains the DC output level trimmer (the Tobey and Dinsdale design doesn't place it here).

This does however increase the VAS current as the HT voltage (offload) is around 73V where it is 48V in the Proprius, and so there is 50% more collector current. The VAS transistor will run hot due to the increased wattage, but we have the "assurances" in Doug Self's book "Audio Power Amplifier Design" that this is normal.

The revised circuit will be published in due course, but it required the "padding" in T1's emitter illustrated by the resistor "R3" and the bypass capacitor shown in the link given above.

HF Stability

Using modern transistors in this vintage configuration exposes its weakness, which is high frequency instability, and it wasn't fully understood, or necessary to be understood, by its early 1960s authors. The power transistors of that day would simply not do the frequencies this design will do. Here they extend beyond 10MHz.

Soon after the publication of this configuration attention turned to the "singleton" input stage referred to very much earlier in this topic. It can be seen that the "singleton" is a current feedback amplifier, and the authors must have seen their configuration as being much the same.

Theirs therefore has frequency compensation to T1's emitter, and in the Leak Stereo 30 (an almost direct copy) "advantage" of the "inherent stability" of current feedback was taken by not having any compensation at all (thanks to Rod Elliott of ESP - http://sound.whsites.net - for pointing these things out).

Simulating zero compensation in this circuit however, it can be seen that even though phase remains well within 90 degrees, the rate of fall suggests oscillation, and is preceded by a rather large spike.

In practice the circuit is perfectly stable as long as signal is never applied! With only a few millivolts of stimulus the current limiting resistor sensibly placed where the HT fuse would go, emitted smoke!

This circuit is therefore not simply a current feedback design. True, it is current feedback by definition, because NFB is applied to the input transistor's emitter, but that is also a route through to T2's base, and could be conceived as being voltage feedback when it is realised that T1 is not a current source for T2's base, as conventional thinking would have it.

It is best seen as HC Lin's (RCA) original circuit, or even better as Mitsubishi's Teleton SAQ300, where both have a single transistor "preamp" before the power amp "for real" and use voltage feedback compensation. In the Teleton the NFB is extended to incorporate the "preamp", and so is current feedback of sorts, but definitely not at DC. Both circuits use the same 470pF value.

The Tobey & Dinsdale uses current feedback compensation, but the only real difference between this and the RCA/Mitsubishi circuits is that T1 is DC coupled. And as that has no bearing on high frequencies it can be seen that they are all really the same.

So we have voltage feedback compensation; current feedback compensation; and no compensation at all if we include the Leak Stereo 30. Who is right?

Obviously the answer is none of them. These were empirical solutions which were the "glue" that held each particular circuit stable - which do not apply universally.

Neither does the global NFB compensation offered by John Linsley Hood. By his own argument of delays, the transistors will have already gone unstable by the time the global compensation "stabilises" them. Simulate it and you will see.

The only conclusion we can draw is that we have both current and voltage feedback at the same time in this circuit configuration. And that this wasn't grasped all those years ago through the lack of simulation software - and the computer operating systems required to run it - and also the lack of high speed transistors such as are being used here.

Settling on 47pf for the voltage feedback (Cdom, T2 collector to base), and 22pf for the current feedback (T2 collector to T1 emitter), with both applied before the delays (phase shifts) of the output stage - and with the output zobel network of the values chosen - we get approximately 90 degrees phase margin, and better than 10dB (12.5dB actually) gain margin.

Remembering that 45 degrees phase margin and 10dB gain margin is stable, we have bettered that. But by obtaining 90 degrees phase margin there should now be no ringing.

By using 47pF T2 collector to base, with the 2mA standing current of T1's collector, we can argue that slew rate is 2/0.047 = 42.5V/uS. Input slew rate calculates as 33V/uS. And with the input filter (1k and 220pf) driven by a zero output impedance source, we have cut from 773kHz. It should not sound distorted at all.

Measurably it is 0.02% THD at 1kHz to 0.1% THD at 20kHz so there is no audible distortion there.

We are however AC coupled at the output, and using 4700uF the capacitor should not introduce much if any low frequency distortion. And little registers on the distortion sweep. But there is one other source which is bridge rectifier RF, which still requires snubbing, but that will have to wait for the new PCBs which have the facility to mount individual rectifier bypass capacitors.


Edited by Graham Slee - Yesterday at 6:57am
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